System and method of providing communications in a wireless power transfer system

ABSTRACT

A communication system that uses keyed modulation to encode fixed frequency communications on a variable frequency power transmission signal in which a single communication bit is represented by a plurality of modulations. To provide a fixed communication rate, the number of modulations associated with each bit is dynamic varying as a function of the ratio of the communication frequency to the carrier signal frequency. In one embodiment, the present invention provides dynamic phase-shift-keyed modulation in which communications are generated by toggling a load at a rate that is a fraction of the power transfer frequency. In another embodiment, the present invention provides communication by toggling a load in the communication transmitter at a rate that is phase locked and at a harmonic of the power transfer frequency. In yet another embodiment, the present invention provides frequency-shift-keyed modulation, including, for example, modulation at one of two different frequencies.

BACKGROUND OF THE INVENTION

The present invention relates to wireless power transfer systems, andmore particularly to systems and methods for providing communications ina wireless power transfer system.

Many conventional wireless power supply systems rely on inductive powertransfer to convey electrical power without wires. A typical inductivepower transfer system includes an inductive power supply that uses aprimary coil to wirelessly transfer energy in the form of a varyingelectromagnetic field and a remote device that uses a secondary coil toconvert the energy in the electromagnetic field into electrical power.Recognizing the potential benefits, some developers have focused onproducing wireless power supply systems with adaptive control systemscapable of adapting to maximize efficiency and provide appropriateoperation to a variety of different types of devices under a wide rangeof circumstances. Adaptive control systems may vary operating parameterssuch as resonant frequency, operating frequency, rail voltage or dutycycle, to supply the appropriate amount of power and to adjust tovarious operating conditions. For example, it may be desirable to varythe operating parameters of the wireless power supply based on thenumber of electronic device(s), the general power requirements of theelectronic device(s) and the instantaneous power needs of the electronicdevice(s). As another example, the distance, location and orientation ofthe electronic device(s) with respect to the primary coil may affect theefficiency of the power transfer, and variations in operating parametersmay be used to optimize operation. In a further example, the presence ofparasitic metal in range of the wireless power supply may affectperformance or present other undesirable issues. The adaptive controlsystem may respond to the presence of parasitic metal by adjustingoperating parameters or shutting down the power supply. In addition tothese examples, those skilled in the field will recognize additionalbenefits from the use of an adaptive control system.

To provide improved efficiency and other benefits, it is not uncommonfor conventional wireless power supply systems to incorporate acommunication system that allows the remote device to communicate withthe power supply. In some cases, the communication system allows one-waycommunication from the remote device to the power supply. In othercases, the system provides bi-directional communications that allowcommunication to flow in both directions. For example, the wirelesspower supply and the remote device may perform a handshake or otherwisecommunicate to establish that the remote device is compatible with thewireless power supply. The remote device may also communicate itsgeneral power requirements, as well as information representative of theamount of power it is receiving from the wireless power supply. Thisinformation may allow the wireless power supply to adjust its operatingparameters to supply the appropriate amount of power at optimumefficiency. These and other benefits may result from the existence of acommunication channel from the remote device to the wireless powersupply.

An efficient and effective method for providing communication in awireless power supply that transfers power using an inductive field isto overlay the communications on the inductive field. This allowscommunication without the need to add a separate wireless communicationlink. One common method for embedding communications in the inductivefield is referred to as “backscatter modulation.” Backscatter modulationrelies on the principle that the impedance of the remote device isconveyed back to the power supply through reflected impedance. Withbackscatter modulation, the impedance of the remote device isselectively varied to create a data stream (e.g. a bit stream) that isconveyed to power supply by reflected impedance. For example, theimpedance may be modulated by selectively applying a load resistor tothe secondary circuit. The power supply monitors a characteristic of thepower in the tank circuit that is impacted by the reflected impedance.For example, the power supply may monitor the current in the tankcircuit for fluctuations that represent a data stream.

A variety of schemes have been developed for encoding data that istransmitted on an inductive field using backscatter modulation. Onecommon approach is bi-phase modulation. Bi-phase modulation uses ascheme in which the signal varies from high to low at every clock pulse.To encode a “1,” the modulator adds an additional transition during thetime period associated with that bit. To encode a “0,” the clock pulsetransition is the only transition to occur during the time periodassociated with that bit.

Wireless power communications can be disrupted if the device beingpowered presents a noisy load. For example, changes in the amount ofpower consumed in a device may change the impedance of the remotedevice. These changes in impedance may create the appearance of datawhen none exists or they may corrupt legitimate data. The power supplycan be especially susceptible to noise that occurs at that samefrequency as the data communications. For example, it is possible thatload fluctuations occurring while data is being transmitted will maskthe legitimate data. As another example, if occurring in the samefrequency range as the data communications, it is possible that a randompattern in the noise will be misinterpreted as the preamble or startbits in a legitimate communication signal. If this occurs, the powersupply may think it is receiving legitimate data and attempt to extractdata, for example, in the form of a data packet, following the fauxpreamble. Although the power supply should eventually determine that thedata packet is not legitimate, the power supply may be occupied with theillegitimate packet, which would delay its ability to recognizelegitimate data.

Further, in some applications, the remote device is configured to send“keep-alive” signals to the wireless power supply. The keep-alive signalmay, for example, tell the wireless power supply that a compatibleremote device that needs power is present. If noise prevents aconsecutive number of keep-alive signals from being recognized by thewireless power supply, the supply of power to the device may bediscontinued. This can be particularly problematic when the remotedevice battery is depleted.

SUMMARY OF THE INVENTION

The present invention provides a communication system that uses keyedmodulation to encode fixed frequency communications on a variablefrequency power transmission signal. In one embodiment, a singlecommunication bit (e.g. a single logic high or logic low) is representedby a plurality of modulations. To provide a fixed communication rate,the number of modulations associated with each bit is dynamicallyvarying as a function of the ratio of the communication frequency to thecarrier signal frequency.

In one embodiment, the present invention provides dynamicphase-shift-keyed modulation. In this embodiment, the present inventionprovides communication by toggling a load in the communicationtransmitter at a rate that is a fraction of the power transferfrequency. For example, the load may be modulated at a frequency that isone-half the power transfer frequency. The communication transmitter maybe configured to modulate on every other waveform, increasing themagnitude of every other waveform. Data is encoded by varying whetherthe modulation takes place on every even waveform or every odd waveform.In one embodiment, the communication transmitter includes a modulationclock operating at a frequency that is ½ the frequency of the carrier.In this embodiment, the output of the modulation clock may be “XOR”edwith the data signal to produce the modulation control signal. The datasignal may have a fixed frequency. In this embodiment, the communicationreceiver may decode the communication signal by timeslicing the coilcurrent (which will correspond with the modulated carrier waveform) andlooking for a DC offset between the two time slices.

In another embodiment, the present invention provides communication bytoggling a load in the communication transmitter at a rate that is phaselocked and at a harmonic frequency of the power transfer frequency. Forexample, the load may be toggled at a frequency that is four times thecarrier frequency. As the load modulation frequency varies withvariations in the carrier frequency, the number of modulations thatoccur during the fixed communication frequency will vary. Data isencoded by varying the modulation applied to the positive and negativehalves of each cycle. During the positive half of the waveform, themodulation signal is generated by “XOR”ing the modulation clock signalwith the data signal. During the negative half of the waveform, themodulation signal is generated by “XOR#”ing (also known as “XNOR”ing)the modulation clock signal and the data signal (i.e. inverse of“XOR”ing the modulation clock signal and the data signal). In thisembodiment, the communication receiver may decode the communicationsignal by producing a buffer copy and an inverted copy of the coilcurrent signal, and then alternately passing time slices of the buffercopy or the inverted copy to the controller. The time slices aresynchronized with the modulation frequency. The controller recognizes ahigh or low signal by looking for a DC offset. For example, a low signalmay result in a negative offset while a high signal may result in apositive offset.

In yet another embodiment, the present invention providesfrequency-shift-keyed modulation. In this embodiment, the communicationtransmitter may be configured to modulate at one of two differentfrequencies. A high signal is encoded by modulating at a first frequencyand a low signal is encoded by modulating at a second frequency. Thefirst frequency may be a fraction, such as ⅛^(th), of the carrierfrequency and the second frequency may be a different fraction, such as1/10^(th), of the carrier frequency. In this embodiment, thecommunication receiver may decode the communication signal by filteringthe coil current and passing it to a frequency discriminator.

The present invention provides simple and effective systems and methodsfor transmitting communications at a fixed frequency using a variablefrequency carrier signal. The systems and methods of the presentinvention provide improved reliability when transmitting communicationsover an inductive field by backscatter modulation. By using a pluralityof modulations for each bit, variations or other defects in one or moremodulations may be averaged out over the plurality of modulations andmay not corrupt the data. Further, communication modulation occursduring both high and low signals so communications do not result indramatic variations in load between high and low signals. In someapplications, timeslicing is used so that the base drive waveformcancels itself out, thereby providing a potentially higher signal tonoise ratio.

These and other objects, advantages, and features of the invention willbe more fully understood and appreciated by reference to the descriptionof the current embodiment and the drawings.

Before the embodiments of the invention are explained in detail, it isto be understood that the invention is not limited to the details ofoperation or to the details of construction and the arrangement of thecomponents set forth in the following description or illustrated in thedrawings. The invention may be implemented in various other embodimentsand of being practiced or being carried out in alternative ways notexpressly disclosed herein. Also, it is to be understood that thephraseology and terminology used herein are for the purpose ofdescription and should not be regarded as limiting. The use of“including” and “comprising” and variations thereof is meant toencompass the items listed thereafter and equivalents thereof as well asadditional items and equivalents thereof. Further, enumeration may beused in the description of various embodiments. Unless otherwiseexpressly stated, the use of enumeration should not be construed aslimiting the invention to any specific order or number of components.Nor should the use of enumeration be construed as excluding from thescope of the invention any additional steps or components that might becombined with or into the enumerated steps or components.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic representation of a wireless power transfer systemin accordance with an embodiment of the present invention.

FIG. 2 is a schematic representation of an embodiment of the wirelesspower transfer system of FIG. 1.

FIG. 3 is a schematic representation of a detector circuit.

FIG. 4 is a simplified circuit diagram of one embodiment of a remotedevice.

FIG. 5 is a simplified circuit diagram of a second embodiment of aremote device.

FIG. 6 is a simplified circuit diagram of a third embodiment of a remotedevice.

FIG. 7 is a simplified circuit diagram of a fourth embodiment of aremote device.

FIG. 8 is a simplified circuit diagram of a fifth embodiment of a remotedevice.

FIG. 9 is a simplified circuit diagram of a sixth embodiment of a remotedevice.

FIGS. 10A through 10H are a series of waveform diagrams representativeof a first communication method.

FIGS. 11A and 11B are simplified circuit diagrams showing a portion of afirst detector circuit.

FIGS. 12A through 12I are a series of waveform diagrams representativeof a second communication method.

FIGS. 13A and 13B are simplified circuit diagrams showing a portion of asecond detector circuit.

FIGS. 14A through 14H are a series of waveform diagrams representativeof a third communication method.

FIG. 15 is a simplified circuit diagram showing a portion of a thirddetector circuit.

FIGS. 16A through 16H are a series of waveform diagrams representativeof a fourth communication method.

FIG. 17 is a plot of signal strength vs. output power for the firstdetector circuit.

FIG. 18 is a plot of signal strength vs. output power for a fourthdetector circuit.

FIGS. 19A and 19B are simplified circuit diagrams showing a portion ofthe fourth detector circuit.

FIGS. 20A and 20B are simplified circuit diagrams showing a portion ofthe fourth detector circuit.

DESCRIPTION OF THE CURRENT EMBODIMENT

I. Overview.

A wireless power transfer system in accordance with an embodiment of thepresent invention is shown in FIG. 1. The wireless power transfer system10 generally includes a wireless power supply 12 and a remote device 14.The wireless power supply 12 is capable of wirelessly supplying power tothe remote device 14 by an inductive field that can be coupled to theremote device 14. The remote device 14 is configured to sendcommunications to the wireless power supply 12 by overlaying thecommunications onto the inductive field produced by the wireless powersupply 12. The wireless power supply 12 generally includes power supplycircuitry 16, signal generating circuitry 18, a power transmitter 20 anda wireless communication receiver 22. The wireless communicationreceiver 22 is configured to receive communication from the remotedevice 14 via backscatter modulation. The remote device 14 generallyincludes a power receiver 24, a communications transmitter 26 and aprinciple load 28. The communication transmitter 26 may include acommunication load 30 that can be selectively applied to vary theimpedance of the remote device 14 and thereby create data signals thatare reflected back to the wireless power supply 12 through reflectedimpedance. The communication transmitter 26 is configured to producesignals by modulating a load a plurality of times during each bit time.The characteristics of each modulation may be independent of thefrequency of the carrier signal and may remain essentially constantregardless of changes in the frequency of the carrier signal. However,the number of modulations occurring during a bit time may be a functionof the frequency of the carrier signal, for example, a fraction or amultiple of the carrier signal frequency. The load may be modulated toprovide an improved form of phase keyed shifting or an improved form offrequency keyed shifting. The communication signals may be decoded bysensing a characteristic of power affected by the modulations andaveraging the sensed characteristic over time.

II. Wireless Power Transfer Systems.

The present invention relates to systems and methods for transmittingcommunications in the context of a wireless power transfer system. Thesystems and methods of the present invention relate to the transfer ofcommunications by overlaying data onto the electromagnetic fields usedto transmit power. The present invention is well-suited for use intransmitting essentially any type of data. For example, thecommunication systems and methods of the present invention may be usedto transmit control signals relating to operation of the wireless powertransfer system, such as signals that identify the remote device,provide wireless power supply control parameters or provide informationin real-time relating to wireless power supply (e.g. current, voltage,temperature, battery condition, charging status and remote devicestatus). As another example, the communication systems and methods maybe used to transfer data unrelated to the wireless power transfersystem, such as transferring information associated with features of theremote device, including synchronizing calendars and to-do lists ortransferring files (e.g. audio, video, image, spreadsheet, database,word processing and application files—just to name a few). The presentinvention is described in the context of various embodiments in whichcommunications are transmitted from the remote device 14 to the wirelesspower supply 12. Although not described in detail, it should beunderstood that the present invention may also be used to transfercommunications from the wireless power supply 12 to the remote device 14(or to a plurality of remote devices).

The present invention is described in connection with an adaptivewireless power supply 12 that is capable of adjusting operatingparameters, such as operating frequency, resonant frequency, railvoltage and/or duty cycle, based on communications (e.g. data) from theremote device 14. Although the present invention is described inconnection with an adaptive wireless power supply, it may be implementedin connection with essentially any type of wireless power supply inwhich the wireless transfer of communications is desired. As notedabove, the wireless power supply 12 of FIG. 1 generally includes powersupply circuitry 16, signal generating circuitry 18, a power transmitter20 and a communications receiver 22. FIG. 2 provides a more detailedschematic of one embodiment of the wireless power supply 12 of FIG. 1.In this embodiment, the power supply circuitry 16 generally includes arectifier 32 and a DC-DC converter 34. The rectifier 32 and DC-DCconverter 34 provide the appropriate DC power for the power supplysignal. The power supply circuitry 16 may alternatively be essentiallyany circuitry capable of transforming input power to the form used bythe signal generating circuitry 18. In this embodiment, the signalgenerating circuitry 18 includes a portion of controller 36 andswitching circuitry 38. The controller 36 is configured, among otherthings, to operate the switching circuitry 38 to apply the desired powersupply signal to the power transmitter 20. In this embodiment, the powertransmitter 20 includes a tank circuit 40 having a primary coil 42 and aballast capacitor 44. In this embodiment, the communication receiver 22includes a detector circuit 46 and portions of controller 36. Thedetector circuit 46 is coupled to the tank circuit 40 in thisembodiment, but may be coupled elsewhere as described in more detailbelow. As can be seen, the wireless power supply 12 of this embodimentincludes a controller 36 that performs various functions, such ascontrolling the timing of the switching circuit 38 and cooperating withthe detector circuit 46 to extract and interpret communications signals.These functions may alternatively be handled by separate controllers orother dedicated circuitry.

The detector circuit described generally above may be implemented in awide variety of different embodiments. For example, the detector circuitmay vary from embodiment to embodiment depending upon the type ofmodulation/demodulation implemented in that embodiment and/or dependingon the details of the power supply circuitry. Further, eachmodulation/demodulation scheme may be implemented using a variety ofdifferent circuits. Generally speaking, the detector circuit isconfigured to produce an output signal as a function of a characteristicof power in the power supply that is affected by data communicatedthrough reflected impedance. For example, with reference to FIG. 3, thedetector circuit 46 may include a sensor 45 to sense the current in thetank circuit 40 and demodulation circuitry 47 to convert the sensedcurrent into a stream of high and low signals in accordance with anembodiment of the present invention. The detector circuit 46 mayalternatively be connected to other points in the power supply 12 wherepower is affected by reflected impedance from the remote device 14. Thedemodulation circuitry 47 may include filtering and conditioningcircuitry (not shown in FIG. 3) to filter and condition the output ofthe sensor. For example, the demodulation circuitry 47 may include bandpass filtering circuitry that functions primarily to attenuate highfrequency oscillations that are above the frequency range of the datacommunications and/or to attenuate low frequency oscillations that arebelow the frequency range of the data communications, including withoutlimitation any DC component in the signal. In some embodiments, thesignal may be passed from filtering and conditioning circuitry to acomparator (not shown in FIG. 3) that converts the signals to a streamof high and low signals. The stream of high and low signals can beprovided to a controller, such as controller 36, which interprets thehigh and low signals as a binary data stream in accordance with anembodiment of the present invention. To facilitate disclosure, specificcircuits will be discussed below in connection with the correspondingcommunications methods.

In the illustrated embodiments, the remote electronic device 14 uses abi-phase encoding scheme to encode data. With this method, a binary 1 isrepresented in the encoded data using two transitions with the firsttransition coinciding with the rising edge of the clock signal and thesecond transition coinciding with the falling edge of the clock signal.A binary 0 is represented by a single transition coinciding with therising edge of the clock signal. Accordingly, the controller isconfigured to decode the comparator output using a corresponding scheme.As described below, the present invention provides a variety ofalternative methods for modulating the biphase encoded data onto a powersignal and for demodulating communications extracted from the powersignal.

A remote device 14 in accordance with an embodiment of the presentinvention will now be described in more detail with respect to FIG. 2.The remote device 14 may include a generally conventional electronicdevice, such as a cell phone, a media player, a handheld radio, acamera, a flashlight or essentially any other portable electronicdevice. The remote device 14 may include an electrical energy storagedevice, such as a battery, capacitor or a super capacitor, or it mayoperate without an electrical energy storage device. The componentsassociated with the principle operation of the remote device 14 (and notassociated with wireless power transfer) are generally conventional andtherefore will not be described in detail. Instead, the componentsassociated with the principle operation of the remote device 14 aregenerally referred to as principle load 28. For example, in the contextof a cell phone, no effort is made to describe the electronic componentsassociated with the cell phone itself.

The remote device 14 generally includes a secondary coil 52, a rectifier54, a communications transmitter 26 and a principle load 28. Thesecondary coil 52 may be a coil of wire or essentially any otherinductor capable of generating electrical power in response to thevarying electromagnetic field generated by the wireless power supply 12.The rectifier 54 converts the AC power into DC power. Although notshown, the device 14 may also include a DC-DC converter in thoseembodiments where conversion is desired. In applications where AC poweris desired in the remote device, the rectifier 54 may not be necessary.The communications transmitter 26 of this embodiment includes acontroller 56 and a communication load 30. In addition to its role incommunications, the controller 56 may be configured to perform a varietyof functions, such as applying the rectified power to the principle load28. In some applications, the principle load 28 may include a powermanagement block capable of managing the supply of power to theelectronics of the remote device 14. For example, a conventionalelectronic device may include an internal battery or other electricalenergy storage device (such as a capacitor or super capacitor). Thepower management block may determine when to use the rectified power tocharge the device's internal battery and when to use the power to powerthe device. It may also be capable of apportioning the power betweenbattery charging and directly powering the device. In some applications,the principle load 28 may not include a power management block. In suchapplications, the controller 56 may be programmed to handle the powermanagement functions or the electronic device 14 may include a separatecontroller for handling power management functions.

With regard to its communication function, the controller 56 includesprogramming that enables the controller 56 to selectively apply thecommunication load 30 to create data communications on the power signalusing a backscatter modulation scheme. In operation, the controller 56may be configured to selectively couple the communication load 30 to thesecondary coil 52 at the appropriate timing to create the desired datatransmissions. The communication load 30 may be a resistor or othercircuit component capable of selectively varying the overall impedanceof the remote device 14. For example, as an alternative to a resistor,the communication load 30 may be a capacitor or an inductor (not shown).Although the illustrated embodiments show a single communication load30, multiple communication loads may be used. For example, the systemmay incorporate a dynamic-load communication system in accordance withan embodiment of U.S. application Ser. No. 12/652,061 entitledCOMMUNICATION ACROSS AN INDUCTIVE LINK WITH A DYNAMIC LOAD, which wasfiled on Jan. 5, 2010, and which is incorporated herein by reference inits entirety. Although the communications load 30 may be a dedicatedcircuit component (e.g. a dedicated resistor, inductor or capacitor),the communication load 30 need not be a dedicated component. Forexample, in some applications, communications may be created by togglingthe principle load 28 or some portion of the principle load 28.

Although shown coupled to the controller 56 in the schematicrepresentation of FIG. 2, the communications load 30 may be located inessentially any position in which it is capable of producing the desiredvariation in the impedance of the remote device 14, such as between thesecondary coil 52 and the rectifier 54.

FIG. 4 shows a circuit diagram of one embodiment of the presentinvention. It should be understood that the circuit diagram is asimplified diagram intended to represent the principle circuitcomponents associated with the creation of communication signals in theremote device 14. In this embodiment, a single load is modulated toprovide communications signals. The FIG. 4 embodiment generally includesa secondary coil 52, full bridge rectifier 54, a load 28, a bulkcapacitor 60 and a communication subcircuit 62. The secondary coil 52may be essentially any inductor, but is a coil of wire in theillustrated embodiment. The rectifier 54 is a full bridge rectifierincluding diodes D1-D4. Alternative rectifier configurations may beused. The load 28 represents the functional load of the remote device14. The bulk capacitor 60 is selected to help smooth and filter powerapplied to load 28. The communication subcircuit 62 may include a loadresistor 30 and a FET 64 connected in series between the load 28 andground. Although not shown in FIG. 4, the controller 56 is operativelycoupled to the FET 64 so that the controller 56 can selectively modulatethe load resistor 30 to generate communication signals.

A variety of alternative communication circuits are shown in FIGS. 5-9.As with FIG. 4, FIGS. 5-9 are simplified circuit diagrams intended toshow the principle circuit components associated with the creation ofcommunication signals in the remote device. FIG. 5 shows an embodimentwith a single communication load 30 that has an independent high siderectifier bridge 66. In this embodiment, a full bridge power rectifier54 is provided to rectify power supplied to the load 28. The powerrectifier 54 includes diodes D1-D4. The communication subcircuit 62includes a communication load 30 and a FET 64 with an independentcommunication bridge 66. The independent communication bridge includesdiodes D5-D6. In operation, the controller 56 (not shown in FIG. 5)operates FET 64 to modulate the communication load 30.

FIG. 6 shows another alternative embodiment in which two separatecommunication subcircuits 62 a-62 b are used to apply the communicationload 30 a-30 b. In this embodiment, a full bridge rectifier 54 isprovided to rectify power applied to the load 28. The full bridgerectifier includes diodes D1-D4. The first communication subcircuit 62 ais connected to the common node of diodes D4-D1 to modulate thecommunication load during one half of the drive waveform. The firstcommunication subcircuit 62 a includes a communication load 30 a and aFET 64 a. The FET 64 a is operatively coupled to the controller 56 (notshown in FIG. 6) so that the controller can selectively modulate thecommunication load 30 a. The second communication subcircuit 62 b isconnected to the common node of diodes D3-D2 to modulate thecommunication load 30 b during the other half of the drive waveform. Inthis embodiment, the second communication subcircuit 62 b is essentiallyidentical to the first communication subcircuit 62 a. The secondcommunication subcircuit 62 b includes a communication load 30 b and aFET 64 b. The FET 64 b is operatively coupled to the controller 56 sothat the controller can selectively modulate the communication load 30b.

The present invention may also be used to modulate a load to applycommunication signals to a split secondary coil (e.g. a center-tappedcoil). For example, FIGS. 7-9 show various alternative remote devicecircuits. FIG. 7 shows a circuit with a single communication subcircuitand a shared full-wave rectifier. In this embodiment, the secondary coil52 is a center-tapped, split coil. The communication subcircuit 62includes a communication load 30 and a FET 64. The FET 64 is operativelycoupled to the controller 56 (not shown in FIG. 7) so that thecontroller 56 can selectively modulate the communication load 30.

FIG. 8 is an alternative embodiment with a center-tapped secondary 52and an independent communication bridge 66. In this embodiment, thepower rectifier 54 includes diodes D1-D2. The communication subcircuit62 is coupled to both sides of the secondary by separate diodes D3-D4.The communication subcircuit 62 includes a communication load 30 and aFET 64 for selectively coupling the communication load 30 to ground. TheFET 64 is operatively coupled to controller 56 so that the controller 56can selectively modulate the communication load 30.

FIG. 9 is yet another alternative embodiment. This embodiment includes asplit secondary 52, a full-wave power rectifier 54, and twocommunication bridges 66 a-66 b with independent communication control.The power rectifier 54 includes diodes D1-D2 arranged between the splitsecondary 52 and the load 28. The first communication bridge 66 a isconnected to the node connecting the first side of the secondary 52 anddiode D1 to modulate the communication load during one half of the drivewaveform. The first communication bridge 66 a includes diode D3 and afirst communication subcircuit 62 a. The first communication subcircuit62 a includes a communication load 30 a and a FET 64 a. The FET 64 a isoperatively coupled to the controller 56 (not shown in FIG. 9) so thatthe controller can selectively modulation the communication load. Thesecond communication bridge 66 b is connected to the node connecting theopposite side of the secondary 52 and diode D2 to modulate thecommunication load 30 b during the other half of the drive waveform. Thesecond communication bridge 66 b includes diode D4 and a secondcommunication subcircuit 62 b. In this embodiment, the secondcommunication subcircuit 62 b is essentially identical to the firstcommunication subcircuit 62 a. The second communication subcircuit 62 bincludes a communication load 30 b and a FET 64 b. The FET 64 b isoperatively coupled to the controller 56 so that the controller canselectively modulation the communication load 30 b.

Although the remote device 14 of FIG. 2 is described with a singlecontroller that handles all of the control functions of the wirelesspower-related components, these functions may be divided across multiplecontrollers. For example, there may be a separate controller to handlecommunications. In applications with separate communication subcircuits,the remote device 14 may include separate controllers for operating theseparate communication subcircuits.

II. Communication Methods.

The present invention provides a variety of alternative communicationmethods that may provide improved performance in the context of wirelesspower transfer systems. These methods may be implemented using thewireless power transfer systems described above or any alternativesystems capable of carrying out the methods of the present invention.For purposes of disclosure, the communication methods of the presentinvention will be described primarily in the context of a wireless powertransfer system incorporating the simplified circuit diagram of FIG. 4.The following paragraphs describe alternative communications methodswith reference to various waveform diagrams. These waveform diagramsinclude a first figure showing the data (i.e. the desired stream of onesand zeros), a second figure showing the data stream encoded usingbiphase modulation, and then a series of figures that show furtherdetails during a short period of time containing a transition from alogic low to a logic high. For purposes of this disclosure, thecommunications methods of the present invention are described inconnection with a carrier waveform operating at 100 kHz, but thefrequency may vary. Although the various communications methods aredescribed in connection with a carrier waveform operating a singlefrequency (e.g. 100 kHz), it should be understood that the frequency ofthe carrier waveform may vary over time and that the communicationsmethods of the present invention will automatically adapt to frequencychanges. In the illustrated embodiments, the carrier waveform frequencymay vary over time between 50 kHz and 200 kHz. For purposes ofdisclosure, the various communication methods are described inconnection with data encoded at a fixed frequency of 2 kHz. This fixedfrequency is merely exemplary and the data encoding frequency may varyfrom application to application.

In one embodiment, the communications are encoded by modulating acommunication load at a rate that is a fraction of the drive frequency,such as an even-integer fraction. For example, in the illustratedembodiment, the communication resistor is modulated at one half thefrequency of the carrier waveform. The modulation signal is created bycombining the modulation clock and the encoded data. More specifically,in this embodiment, the modulation clock waveform is XORed with theencoded data waveform to produce the modulation signal. This methodologywill be described in more detail with references to FIGS. 10A-10H. FIG.10A shows a sample data stream of 1s and 0s. FIG. 10B shows the sampledata stream encoded using a biphase encoding stream. Referring now toFIGS. 10C-10H, the modulation signal is created by combining themodulation clock and the encoded data. FIGS. 10C-10H show a shortsegment of the data stream during which there is a transition from a lowsignal to a high signal. FIG. 10C shows the carrier waveform, which asnoted above is about 100 kHz for this illustration. FIG. 10D shows themodulation clock signal. The frequency of the modulation clock signal isone-half the frequency of the carrier (or about 50 kHz) in thisillustration. The data stream is shown in FIG. 10E. As noted above, thedata signal (FIG. 10D) is XOR with the encoded data signal (FIG. 10E) toproduce the XOR waveform shown in FIG. 10F. As can be seen, themodulation clock waveform is copied when the encoded data is low and isinverted when the encoded data is high. When the modulated signal isapplied to the carrier waveform, the resulting modulated carrierwaveform is shown in FIG. 10G. Alternate time slicing of the resultingmodulated carrier waveform is shown in FIG. 10H.

The communication signal may be received, demodulated and decoded usinga variety of alternative systems and methods. For purposes ofdisclosure, the present invention will be described in connection withcommunication receiver 22 of FIG. 3 and the demodulation circuitry ofFIGS. 11A-B. In operation of this embodiment, the current sensor 45produces a signal that is representative of the current in the tankcircuit 40 (see FIG. 3). The current sensor 45 may be a current sensetransformer that produces a signal having a voltage that varies inproportion with the magnitude of the current in the tank circuit 40. Asanother alternative, the current sensor 45 may be an output taken from adivider having a scaling resistor and capacitor as shown in FIGS. 4-9.As noted above, the current sensor 45 may be replaced by essentially anydetector or similar circuit component capable of producing a signal thatis representative of a characteristic of power in the power supply 12that is affected by the reflected impedance of the remote device 14.

In this embodiment, the detector circuit 46 includes a pair ofamplifiers 102 a-102 b that produce a buffer copy and an inverted buffercopy of the signal output by the current sensor (see FIGS. 11A-B). Asshown, in this embodiment, the current sensor signal may be passed toboth an amplifier and an inverting amplifier arranged in parallel withrespect to one another. The output of the amplifier and the invertingamplifier may be passed to a pair of multiplexors 104 a-104 b that arecoupled to a time slicing clock that is synchronized with the modulationclock in the remote device 14. The time slicing clock controls whetherthe buffer copy or the inverted copy of the current signal is passed tothe remainder of the detector circuitry. In this embodiment, the clocksignal is synchronized to the drive frequency divided by two. As shown,in this embodiment, the two multiplexors 104 a-104 b include oppositeNO/NC inputs to provide a differential signal. As an alternative toflipping the NO/NC inputs, the multiplexor clocking may be inverted toprovide a differential signal. In some cases, it may be desirable toclock the multiplexors 104 a-104 b with a signal that is a quadraturecopy of the drive signal. The 90 degree phase shift may allow thecircuitry to better capture the signal. Although the multiplexors of theillustrated embodiment have two inputs, the multiplexors 104 a-104 bcould alternatively have a single input and the output may be leftfloating on the alternate clock phase. This could reduce the signalstrength of the amplifier chain. In this embodiment, the clock signalmay be derived from a variety of sources, such as the drive signal, theprimary coil voltage, the primary coil current or a 90 degree shiftedversion of any of the foregoing.

Referring again to FIGS. 11A-B, multiplexor 104 a passes the buffer copyduring the “A” time slices and passes the inverted copy during the “B”time slices, and multiplexor 104 b passes the buffer copy during the “B”time slices and passes the inverted copy during the “A” time slices.

In the detector circuit of FIGS. 11A-B, the output of each multiplexor104 a-104 b is passed through a separate amplifier chain. In theillustrated embodiment, the output of each multiplexor 104 a-104 b ispassed to separate averaging circuitry 106 a-106 b. Each of theseaveraging circuits 106 a-106 b outputs the average of the minimum andmaximum of its respective input, which may provide improved performanceover a straight average in some applications because it may be lessinfluenced by the shape of the waveform and more sensitive to extremevalues. Although potentially beneficial in some applications, theoutputs need not be averages of the minimum and maximum. For example, insome applications, the averaging circuits 106 a-106 b may alternativelyoutput a straight average of their respective input signals.

In the illustrated embodiment, the outputs of the averaging circuits 106a-106 b are passed to separate low pass filters 108 a-108 b. In thisembodiment, the filters 108 a-108 b may be two pole 5 kHz low passfilters. These low pass filters 108 a-108 b function primarily to removethe AC components of the signal above the communication frequency range.Although this function is performed in the illustrated embodiment withop-amps, the op-amps may be replaced by alternative filtering circuitry,such as a passive filter or a digital filter.

In some applications, it may be desirable to amplify the outputs of thelow pass filters 108 a-108 b. In the illustrated embodiment, the outputsof the low pass filters 108 a-108 b are passed to separate amplifiers110 a-110 b. In the illustrated embodiment, the amplifiers 110 a-110 bare AC coupled amplifiers that amplify the filtered signal, maintaininga center point around Vbias. In this embodiment, the AC coupling removesany DC offset and serves as a single pole high pass filter.

The outputs of the amplifiers 110 a-110 b are passed to separate lowpass filters 112 a-112 b. These low pass filters 112 a-112 b remove ACcomponents of the signal above the communication frequency range andhelp to remove noise imparted by the AC amplifier 110 a-110 b. Althoughlow pass filters 112 a-112 b are implemented in the illustratedembodiment with op-amps, the op-amps may be replaced by alternativefiltering circuitry, such as a passive filter or a digital filter. Insome applications, the signal-to-noise ratio of the outputs of theamplifiers may be sufficient so that low pass filters 112 a-112 b areunnecessary.

In the illustrated embodiment, the outputs of the final low pass filters112 a-112 b are separately passed to a comparator 114. The comparator114 combines the differential signals from the two amplifier chains backinto a single, “digitized” signal that can be readily decoded by amicrocontroller, such as controller 36. Referring to FIG. 10G, whenlogic low is being sent, time slices “A” have a larger negative peak andtime slices “B” have a larger positive peak. This condition results in alow amplifier output. Conversely, when logic high is being sent, timeslices “A” have a larger positive peak and time slices “B” have a largernegative peak. This condition results in a high amplifier output. Thestream of high and low signals output by the comparator may be decodedusing the same scheme used to encode the data in the remote device 14,which in this embodiment is a biphase encoding scheme.

As an alternative to the dual-chain circuitry of FIGS. 11A-B, thedetector circuit 46 may alternatively incorporate a single endeddetection chain. In such alternatives, the detector circuit may includeonly the topmost chain of FIGS. 11A-B. The dual-chain circuitry of FIGS.11A-B may provide improved performance in some applications because thedifferential pair of amplifier chains provides improved signal-to-noiseratio as one signal is increasing in voltage while the other isdecreasing. As a result, DC drift is unlikely to distort the signal. Thecomparator of this alternative embodiment is configured to provide ahigh or low output based on a comparison of the amplitude of the inputsignal with a reference signal. More specifically, the comparator mayinclude a first input that receives the signal from the single detectionchain and a second input that is coupled to a reference signal, Vbias.The reference signal may be set to be slightly lower than the DCcomponent of the amplified signal. Accordingly, the comparator outputwill remain high when no communication is present in the signal. Ifcommunication is present, then the comparator output toggles betweenhigh and low in correspondence with the communication signals. Theoutput of the comparator 114 is passed to the controller 36 fordecoding. In this case, decoding is achieved using a biphase decodingscheme.

In an alternative embodiment, communications are encoded by modulating acommunication load at a rate that is a fraction of the drive frequency,similar to the embodiment described above with respect to FIGS. 10A-Hand 11A-B, with some exceptions. Rather than modulating thecommunication load during a full cycle of the drive waveform, shown forexample in FIG. 10F, the communication load in this embodiment ismodulated during a fraction of the drive waveform, such as one-half thedrive waveform.

Referring to FIGS. 16A-H, in the illustrated embodiment, a communicationload is modulated at one-half the frequency of the carrier waveform. Themodulation clock waveform may be XORed with the encoded data waveform toproduce a modulator control waveform. FIGS. 16C-H show a short segmentof the data stream during which there is a transition from a low signalto a high signal. FIG. 16C shows the carrier waveform, which as notedabove is about 100 kHz for this illustration. FIG. 16D shows themodulation clock waveform having a frequency that is one-half thefrequency of the carrier (or about 50 kHz) in this illustration. Thedata stream is shown in FIG. 16E, which after being XORed with themodulation waveform yields the XOR waveform of FIG. 16F. It should beappreciated that up to this point the features of this alternativeembodiment are substantially similar to the embodiment described withrespect to FIGS. 10A-H.

As described in the previous embodiment, the communication load may bemodulated according to the XORed waveform for a full cycle of thecarrier waveform, or in other words, at a 50% duty cycle using amodulated clock waveform having a frequency that is one-half the carrierfrequency. However, in this alternative embodiment, the XORed waveformis applied for approximately one-half the cycle of the carrier waveform,or in other words, at a 25% duty cycle using a modulated clock waveformhaving a frequency that is one-half the carrier frequency. Thus, thecommunication load may be applied for less time than in the previousembodiment, and to increase the magnitude of either a peak or atough—but not both—of every other waveform of the carrier waveform inorder to communicate the data.

This 25% duty cycle modulation may be achieved by generating an XORwaveform similar to the XOR waveform illustrated in the embodiments ofFIGS. 10A-H; but rather than applying the XOR waveform during a fullcycle of the carrier waveform, the XOR waveform may be applied duringone-half of each cycle of the carrier waveform. For example, using thecommunication circuits of FIGS. 6 and 9, applying the XOR waveform forone-half of each cycle of the carrier waveform may be accomplished byselectively modulating one of the communication loads 30 a-30 b—notboth—or put differently, modulating one leg of the secondary coil 52instead of both legs. It should be appreciated that although thisembodiment implements a 25% duty cycle using an XOR waveform similar tothat of the embodiments above, the 25% duty cycle may be achievedinstead by modifying the modulation clock waveform to have a 25% dutycycle and XORing it with the encoded data waveform. Using thecommunication circuits of FIGS. 4-5 and 7-8, for example, thecommunication load 30 may be modulated according to this XORed waveformin order to apply the communication load 30 for approximately one-halfof each cycle of the carrier waveform.

Propagation delays inherent to the electronics of this embodiment, andother embodiments, may cause the modulation clock to be delayed withrespect to the carrier waveform. In the illustrated embodiment of FIGS.16C-D, this propagation delay can be seen by the slight shift in themodulation clock waveform with respect to the zero crossing of thecarrier waveform. In some embodiments, for example, this propagationdelay may affect when the communication load is applied relative to thecarrier waveform, and whether the remote device 14 attempts to adapt orcompensate for the delay.

In the illustrated embodiments of FIGS. 16A-H, the propagation delay mayaffect the choice of which leg of the secondary coil 52 to modulate inorder to implement 25% duty cycle modulation. Using the circuit topologyof FIG. 6 as a reference, the current through the load resistor 30 a(coupled to the first leg of the secondary coil 52) during modulationcycles is shown in FIG. 16H, and the current through the load resistor30 b (coupled to the second leg of the secondary coil 52) duringmodulation cycles is shown in FIG. 16G. Because the modulation clockwaveform of FIG. 16D is synced with the carrier waveform at the firstleg of the secondary coil 52, a propagation delay prevents fullmodulation of the load resistor 30 a as illustrated by the choppedmodulator current in FIG. 16H. As a result, in order to account for theeffects of propagation delay and achieve full modulation at 25% dutycycle, a communication load may be modulated on a leg of the secondarythat is opposite the leg which generates the modulation clock waveform.For example, in the illustrated embodiment of FIG. 16G, full modulationis achieved at 25% duty cycle by modulating the load resistor 30 bcoupled to the second leg of the secondary coil 52.

Similar to other alternative embodiments discussed herein, thecommunication load of this embodiment may be one or more impedanceelements, such as a load resistor or one or more alternative components.For example, the communication load may be resistive, capacitive, orinductive, or a combination thereof. Although FIGS. 4-9 show a loadresistor 30, this embodiment may in some cases function better with aload capacitor in place of the load resistor 30.

The two plots of FIGS. 17 and 18 illustrate potential differencesbetween embodiments of FIGS. 16A-H that modulate at 25% duty cycle usinga single load resistor 30 a and a 50% duty cycle modulator. FIG. 17shows communication nulls or inversion that may result from depletingthe energy in the coils for two consecutive cycles with embodimentsusing a 50% duty cycle modulator. Specifically, at output powers ofapproximately 15-30 W, the modulation depth may invert or go below 0.0mV, causing a null communication zone. As shown in FIG. 18, with a 25%duty cycle modulator applying a communication load to one side of thecoil, the signal strength decreases in some zones due to the modulation,but inversion and null communication zones are avoided. In one aspect,the present invention may provide communication circuitry that avoidscommunication inversion. Further potential attributes of the 25% dutycycle embodiments may be a reduction or elimination of audible noiseduring communication, though 50% duty cycle embodiments may also reduceor eliminate audible noise.

Referring now to FIGS. 19A-B and 20A-B, the communication signal may bereceived, demodulated, and decoded using systems and methods similar tothe embodiments described above with respect to FIGS. 11A-B, with someexceptions. Instead of averaging circuitry 106 a-b, this embodiment usespeak detector circuitry 106 a-b″ and 106 a-b′″ and includes two detectorcircuits 46″ and 46′″ to demodulate the communication signal. Otherwise,this embodiment and its alternatives are similar to other embodimentsdescribed herein.

As discussed above, in embodiments that utilize 25% duty cyclemodulation, the communication load may be applied during one-half of thecarrier wave cycle. As a result, the modulation may cause a level shiftto be reflected through the inductive coupling to the wireless powersupply that generally affects either the peaks or the troughs—but notboth—of the current sensed by the current sensor. In other words, with25% duty cycle modulation, (a) levels of the peak current or voltagethrough the primary coil from even to odd cycles may shift, or (b)levels of the trough current or voltage through the primary coil fromeven to odd cycles may shift. If the dot orientation (e.g. windingorientation) of the primary coil 42 with respect to the secondary coil52 of the remote device is unknown, it may not be known whether thelevel shift occurs in the trough or the peak. Accordingly, thisembodiment utilizes two detector circuits 46″ and 46′″ to sense levelshifts in either the peaks or the troughs due to communication loadmodulation. The respective outputs (A and B) of the two detectorcircuits 46″ and 46′″ are then combined, such as being ORed together, toyield a digital representation of the communication signal modulatedthrough the inductive coupling. In alternative embodiments in which thedot orientation is known, a single detector circuit may be used todetect level shifts in either the peaks or the troughs, whichever isexpected, depending on the known dot convention and which leg of thesecondary 52 is being modulated. It should be appreciated that knowingthe dot orientation may be unnecessary in embodiments that use 50% dutycycle modulation because level shifts due to modulation of thecommunication load occur in both the peaks and the troughs.

The first and second detector circuits 46″ and 46′″ are now described infurther detail. The first detector circuit 46″ produces a buffer copy ofthe signal output by the current sensor, and a second detector circuit46′″ produces an inverted buffer copy of the signal output by thecurrent sensor. In this way, the first detector circuit 46″ samples andtime slices peaks of the carrier wave to detect a level shift inresponse to application of a communication load, and the second detectorcircuit 46′″ samples and time slices the troughs of the carrier wave todetect a level shift in response to application of a communication load.

Referring again to FIGS. 16A-H along with FIGS. 19A-B, the multiplexor104 a″ of detector 46″ passes a buffered copy during “A” time slices,and the multiplexor 104 b″ of detector 46″ passes a buffered copy during“B” time slices. Accordingly, the “A” time slices and the “B” timeslices are passed through a separate amplifier chain, which with someexceptions may be similar to the amplifier chain of the illustratedembodiment of FIGS. 11A-B. In the illustrated embodiment of FIGS. 19A-B,the outputs of the multiplexors 104 a-b″ are passed to separate peakdetectors 106 a-b″. Each of the peak detectors 106 a-b″ outputs the peakvalue of its respective input during a time slice. This peak detectionmay provide improved signal detection by capturing level shiftsresulting from modulation of the communication load. The peak detectormay also cancel out any asymmetry imposed by half-bridge driver hardwareof the wireless power supply. For purposes of disclosure, the peakdetector 106 a-b″ is used in the illustrated embodiment of FIGS. 19A-B,but the peak detector may also be implemented in other embodimentsdescribed herein. In further alternatives, the peak detector may bereplaced by a trough detector, which may detect minimum conditionsrather than maximum conditions in the signal.

The outputs of the peak detectors 106 a-b″ are each passed throughseparate amplifier chains as discussed above, and eventually comparedagainst each other using a differential amplifier, such as comparator114″. If a level shift is detected between the peaks of the buffered,non-inverted signal in the “A” time slices as compared to the “B” timeslices, the comparator 114″ outputs a “digitized” signal that can bereadily decoded by a microcontroller, such as controller 36. As anexample, if the peak value of the signal during the “A” time slices ishigher than the peak value of the signal during the “B” time slices, alogic high will be output from the detector 46″. Conversely, if the peakvalue of the signal during “A” time slices is lower than the peak valueof the signal during the “B” time slices, then a logic low will beoutput from the detector 46″.

Turning to the illustrated embodiment of FIGS. 20A-B, the detector 46′″detects the opposite of detector 46″; rather than detecting peaks in thesignal, detector 46′″ detects troughs or minimums in the signal. Forexample, by detecting a level shift in the troughs, detector 46′″ maydemodulate data encoded within the signal.

In particular, the multiplexor 104 a′″ of detector 46′″ passes abuffered inverted copy of the signal during “A” time slices, and themultiplexor 104 b″ passes a buffered inverted copy of the signal during“B” time slices. Accordingly, the “A” time slices and the “B” timeslices are passed through a separate amplifier chain, which with someexceptions is similar to the amplifier chain of the illustratedembodiment of FIGS. 11A-B. In the illustrated embodiment of FIGS. 20A-B,the outputs of the multiplexors 104 a-b″ are passed to separate peakdetectors 106 a-b″, which in detector 46′″ detect troughs or minimums inthe signal. Each of the peak detectors 106 a-b′″ outputs the peak valueof the inverted signal (the trough of the non-inverted signal) of itsrespective input during a time slice.

Similar to detector 46″, the outputs of the peak detectors 106 a-b′″ areeach passed through separate amplifier chains, and eventually comparedagainst each other using a differential amplifier, such as comparator114″. As before, if a level shift is detected between the troughs of thebuffered, non-inverted signal in the “A” time slices as compared to the“B” time slices, the comparator 114′″ outputs a “digitized” signal thatcan be readily decoded by a microcontroller.

As described above, without knowing the dot orientation of the primarywith respect to the secondary, it may be uncertain whether modulation ofthe communication load at 25% duty cycle will effect a level shift inthe troughs or the peaks of the signal. As a result, the respectiveoutputs (A and B) of both detector 46″ and detector 46′″ may bemonitored by a microcontroller so that the signal can be detected andreadily decoded.

As an example, using 25% duty cycle modulation by modulating the secondleg of the secondary coil (FIG. 16G), operation of the detectors 46″ and46′″ would yield the following results. It should be appreciated thatresults may differ depending on the manner of modulating thecommunication load. In the example embodiment of FIGS. 16A-G, as appliedto the illustrated embodiment of FIGS. 19A-B and 20A-B, detector 46″would not detect a level shift between peak values of the buffered,non-inverted signal during the “A” time slices and the “B” time slices.As shown in FIG. 16G, the communication load is being modulated duringthe trough cycle of the carrier wave, not the peak cycle, and thereforethe peaks of the signal are generally unaffected by modulation of thecommunication load.

Detector 46″, on the other hand, would detect a level shift between thepeak values of the buffered, inverted signal during the “A” time slicesand the “B” time slices. Because the communication load is beingmodulated during the trough cycle of the carrier wave, the peaks of theinverted form of the signal (troughs of the non-inverted signal), asmeasured by peak detectors 106 a-b′″ would identify a level shiftbetween the “A” time slices and the “B” time slices, and output thelevel shift as a “digitized” signal to be decoded by a microcontrolleras described previously.

In another alternative embodiment, communications are encoded bymodulating a communication load at a rate that is a multiple of thedrive frequency, such as an even-integer multiple. For example, in theillustrated embodiment, the communication resistor is modulated at fourtimes the frequency of the carrier waveform. This embodiment may includea phase lock loop (“PLL”) to generate a modulation clock waveform thatremains in phase with the carrier waveform. FIG. 12A shows a sample datastream of 1s and 0s. FIG. 12B shows the sample data stream encoded usinga biphase encoding stream. Referring now to FIGS. 12C-12I, themodulation signal is created by combining the modulation clock and theencoded data using different functions for the positive and negativehalves of each cycle of the waveform. More specifically, in thisembodiment, during the positive half of the carrier waveform, themodulation clock waveform is XORed with the encoded data waveform, andduring the negative half of the carrier waveform, the modulation clockwaveform is XOR#ed with the encoded data waveform. This may simplifydemodulation by the wireless power supply. FIG. 12C shows the carrierwaveform, which as noted above is about 100 kHz. FIG. 12D shows themodulation clock signal. In this embodiment, the frequency of themodulation clock signal is four times the frequency of the carrier orabout 400 kHz in this illustration. The data stream is shown in FIG.12E. As noted above, this illustration shows a short portion of the datastream including a single transition for low to high. The XOR waveformis shown in FIG. 12F and the XOR# waveform is shown in FIG. 12G. Thecomposite waveform resulting from alternately XORing or XOR#ing the datasignal with the modulation clock signal provides the modulated signal asshown in FIG. 12H. When the modulated signal is applied to the carrierwaveform, the resulting modulated carrier waveform is shown in FIG. 12I.It should be noted that FIG. 12I represents an idealized waveform. Inpractice, the modulated signal is unlikely to produce instantaneousvariations in the primary current. Instead, the current is likely totake some time to transition and the actual waveform is likely to havetransition regions rather than clean jumps between time slices.

The communication signal produced by this second communications methodmay be received and decoded using a variety of alternative systems andmethods. For purposes of disclosure, the present invention will bedescribed in connection with communication receiver 22 of FIG. 2 and thedemodulation circuitry of FIGS. 13A-B. As will be seen, the demodulationcircuitry of FIGS. 13A-B is similar to the demodulation circuitry ofFIGS. 11A-B. It does not, however, include averaging circuitry 106 a-106b.

In operation of this embodiment, the current sensor 45 produces a signalthat is representative of the current in the tank circuit. The currentsensor 45 may be a current sense transformer that produces a signalhaving a voltage that varies in proportion with the magnitude of thecurrent in the tank circuit 40. As another alternative, the currentsensor 45 may be an output taken from a divider having a scalingresistor and capacitor as shown in FIGS. 4-9. The current sensor 45 maybe replaced by essentially any detector capable of producing a signalthat is representative of a characteristic of power in the power supply12 that is affected by the reflected impedance of the remote device 14.

In this embodiment, the detector circuit 46′ includes a pair ofamplifiers 102 a′-102 b′ that produce a buffer copy and an invertedbuffer copy of the signal output by the current sensor. As shown, inthis embodiment, the current sensor signal may be passed to an amplifierand an inverting amplifier arranged in parallel with respect to oneanother. The output of the amplifier and the inverting amplifier may bepassed to a pair of multiplexors 104 a′-104 b′ that are coupled to atime slicing clock that is synchronized with the modulation clock in theremote device 14. The time slicing clock controls whether the buffercopy or the inverted copy of the current signal is passed to theremainder of the detector circuitry. In this embodiment, the clocksignal is synchronized to the drive frequency multiplied by four. Asshown, in this embodiment, the two multiplexors 104 a′-104 b′ includeopposite NO/NC inputs to provide a differential signal. As analternative to flipping the NO/NC inputs, the clocking of onemultiplexor with respect to another may be inverted to provide adifferential signal. For example, multiplexor 104 b′ clocking may beinverted with respect to multiplexor 104 a′ clocking in order to providea differential signal. In some cases, it may be desirable to clock themultiplexors 104 a′-104 b′ with a signal that is a quadrature copy ofthe drive signal. The 90 degree phase shift may allow the circuitry tobetter capture the signal. Although the multiplexors of the illustratedembodiment have two inputs, the multiplexors 104 a′-104 b′ couldalternatively have a single input and the output may be left floating onthe alternate clock phase. This could reduce the signal strength of theamplifier chain. In this embodiment, the clock signal may be derivedfrom a variety of sources, such as the drive signal, the primary coilvoltage, the primary coil current or a 90 degree shifted version of anyof the foregoing.

Referring again to FIGS. 13A-B, multiplexor 104 a′ passes the buffercopy during the “A” time slices and passes the inverted copy during the“B” time slices, and multiplexor 104 b′ passes the buffer copy duringthe “B” time slices and passes the inverted copy during the “A” timeslices.

In the detector circuit of FIGS. 13A-B, the output of each multiplexor104 a′-104 b′ is passed through a separate amplifier chain. In theillustrated embodiment, the output of each multiplexor 104 a′-104 b′ ispassed to a separate low pass filter 108 a′-108 b′. In this embodiment,the filters 108 a′-108 b′ may be two pole 5 kHz low pass filters. Theselow pass filters 108 a′-108 b′ function primarily to remove the ACcomponents of the signal above the communication frequency range.Although this function is performed in the illustrated embodiment withop-amps, the op-amps may be replaced by alternative filtering circuitry,such as a passive filter or a digital filter.

In some applications, it may be desirable to amplify the outputs of thelow pass filters 108 a′-108 b′. In the illustrated embodiment, theoutputs of the low pass filters 108 a′-108 b′ are passed to separateamplifiers 110 a′-110 b′. In the illustrated embodiment, the amplifiers110 a′-110 b′ are AC coupled amplifiers that amplify the filteredsignal, maintaining a center point around Vbias. In this embodiment, theAC coupling removes any DC offset and serves as a single pole high passfilter.

The outputs of the amplifiers 110 a′-110 b′ are passed to separate lowpass filters 112 a′-112 b′. These low pass filters 112 a′-112 b′ removeAC components of the signal above the communication frequency range andhelp to remove noise imparted by the AC amplifier 110 a′-110 b′.Although low pass filters 112 a′-112 b′ are implemented in theillustrated embodiment with op-amps, the op-amps may be replaced byalternative filtering circuitry, such as a passive filter or a digitalfilter. In some applications, the signal-to-noise ratio of the outputsof the amplifiers may be sufficient so that low pass filters 112 a′-112b′ are unnecessary.

In the illustrated embodiment, the outputs of the final low pass filters112 a′-112 b′ are separately passed to a comparator 114′. The comparator114′ combines the differential signals from the two amplifier chainsback into a single, “digitized” signal that can be readily decoded by amicrocontroller, such as controller 36. As can be seen in FIG. 12I, thecomposite waveform combines to provide a negative DC offset when theencoded data is low, and to provide a positive DC offset when theencoded data is high. The output of the comparator 114′ may be passed tothe communication controller 36 for decoding. In this case, thecommunication controller 36 will decode the binary stream using the samescheme used to encode the data in the remote device 14, which in thisillustration is a biphase encoding scheme.

As an alternative to the dual-chain circuitry of FIGS. 13A-B, thedetector circuit 46′ may alternatively incorporate a single endeddetection chain. In such alternatives, the detector circuit may includeonly the topmost chain of FIGS. 13A-B, and the comparator 114′ may haveits negative input referenced to Vbias. The dual-chain circuitry ofFIGS. 13A-B may provide improved performance in some applicationsbecause the differential pair of amplifier chains provides improvedsignal-to-noise ratio as one signal is increasing in voltage while theother is decreasing. As a result, DC drift is unlikely to distort thesignal.

In yet another alternative embodiment, the communications are encoded bymodulating a communication load at one of two different frequencies. Inone embodiment, the two different frequencies may be fixed frequencies.Having two different fixed frequencies for modulation may simplify thefilter design for demodulation in the wireless power supply. In oneembodiment, the two different frequencies may be fractions or multiplesof the drive frequency, such as an integer fraction or an integermultiple. For example, in the illustrated embodiment, the communicationresistor is modulated at one frequency to represent a logic high and ata different frequency to represent a logic low. In this embodiment, thecommunications controller 36 includes two different modulationclocks—one at ⅛^(th) the frequency of the carrier waveform and the otherat 1/10^(th) the frequency of the carrier waveform. The frequency of themodulation clocks may vary from application to application. Themodulation signal is a composite of the two modulation clocks created bycombining the first modulation clock during a logic low and the secondmodulation clock during a logic high. The frequency shifting methodologyof this embodiment will be described in more detail with references toFIGS. 14A-14H. FIG. 14A shows a sample data stream of 1s and 0s. FIG.14B shows the sample data stream encoded using a biphase encodingscheme. The process for creating the modulated carrier waveform isdescribed in more detail with reference to FIGS. 14C-14H, which show ashort portion of the data stream containing a transition from a lowsignal to a high signal. FIG. 14C shows the carrier waveform, which asnoted above is about 100 kHz for this illustration. FIG. 14D shows themodulation clock A signal. In this illustration, the frequency ofmodulation clock A is one eighth the frequency of the carrier. FIG. 14Eshows the modulation clock B signal. In this illustration, the frequencyof modulation clock B is one tenth the frequency of the carrier. Thedata stream is shown in FIG. 14F. The modulated carrier waveform iscreated by combining the carrier waveform (FIG. 14C) with eithermodulation clock A signal (FIG. 14D) or modulation clock B signal (FIG.14E) using the encoded data signal (FIG. 14F) as a key. Morespecifically, during a low data signal, the carrier waveform is combinedwith the modulation clock A signal and, during a high data signal, thecarrier waveform is combined with the modulation clock B signal. Theresulting modulation clock waveform is shown in FIG. 14G. As can beseen, the modulation clock waveform modulates at one frequency while theencoded data is low, and switches to modulation at the other frequencywhen the encoded data is high. As can be seen, the waveform modulatesmore quickly for a logic low than for a logic high. In thisillustration, each modulation includes four cycles of the carrierwaveform during a logic low and five cycles of the carrier waveformduring a logic high. The frequencies of the carrier waveform and the twomodulation clock signals can be varied to alter the characteristics ofthe modulated carrier waveform.

The communication signal produced by this alternative communicationsmethod may be received and decoded using a variety of alternativesystems and methods. For purposes of disclosure, the present inventionwill be described in connection with communication receiver 22 describedabove and in connection with FIG. 3 and FIG. 15. In operation of thisembodiment, the current sensor 45 produces a signal that isrepresentative of the current in the tank circuit. The current sensor 45may be a current sense transformer that produces a signal having avoltage that varies with the magnitude of the current in the tankcircuit 40. As another alternative, the current sensor 45 may be anoutput taken from a divider having a scaling resistor and capacitor asshown in FIGS. 4-9. The signal from the current sensor 45 may be passedthrough filtering and conditioning circuitry, such as a bandpass filterand an amplifier. The bandpass filter may include a low pass filter 202and a high pass filter 204. The filters may be configured to filter outsignals below the lowest frequency of operation divided by ten and abovethe highest frequency of operation divided by eight. The amplifier 206may be configured to amplify the signal to an appropriate level for thefrequency discriminator (not shown). The amplified output may be passedto the frequency discriminator (not shown). In one embodiment, thefrequency discriminator is an integrated circuit system (ICS) thatconverts frequency to a voltage. The output of the ICS may be providedto controller 36 for decoding. In an alternative embodiment, thefrequency discriminator may include a comparator (not shown) and acontroller, such as controller 36. The comparator may produce a squarewave output and the controller 36 may count the edges to determine themodulation frequency. In embodiments that incorporate a comparator, itmay be desirable to incorporate a filter between the amplifier 206 andthe comparator (not shown) depending in large part on the noise withinthe circuitry. The system may interpret a signal that alternates betweenhigh and low every four cycles to be a logic low, and the system mayinterpret a signal that alternates between high and low every fivecycles to a logic high. In both of these embodiments, the communicationcontroller 36 will decode the binary stream of logic highs and logiclows using a biphase decoding scheme.

Although the present invention is described in connection with adetector circuit 46 having analog circuitry for filtering andconditioning the signal before it is fed to the controller 36, thefiltering, conditioning and/or comparator functions may alternatively becarried out using a digital signal processor (DSP). For example, in onealternative, the output of current sense transformer (or other detector)may be fed into a DSP (not shown). The DSP may convert the analog signalinto a digital signal and then processes the digital signal to generatehigh and low outputs that are consistent with the high and low outputsthat would have been generated using the circuitry described above. TheDSP may process the input signal to remove signal components occurringoutside the frequency range used for communications, analyze theremaining signal to identify the communication signals then provide anoutput signal that drives high and low with the communication signals.

In the illustrated embodiments, the communication receiver includes adetector circuit that demodulates communications through a current sensetransformer that provides an output representative of the current in thetank circuit. The communication receiver may operate using alternativemethods and apparatus. For example, the power supply may include adetector circuit that provides a signal indicative of the current in theinput to the switching circuit (e.g. an amplifier coupled to the inputof the switching circuit). As another example, the power supply mayinclude a detector circuit that detects communications using the phaserelationship of the voltage of the input to the switching circuit andthe current in the tank circuit. As a further example, the power supplymay include a detector circuit that detects communication using thevoltage in the tank circuit. Operation of some of these alternativesystems and methods for detecting communications is described in moredetail in U.S. Provisional Application No. 61/298,021, entitled SYSTEMSAND METHODS FOR DETECTING DATA COMMUNICATION OVER A WIRELESS POWER LINKand filed on Jan. 25, 2010, which is incorporated herein by reference inits entirety.

The above description is that of current embodiments of the invention.Various alterations and changes can be made without departing from thespirit and broader aspects of the invention. This disclosure ispresented for illustrative purposes and should not be interpreted as anexhaustive description of all embodiments of the invention or to limitthe scope of the claims to the specific elements illustrated ordescribed in connection with these embodiments. For example, and withoutlimitation, any individual element(s) of the described invention may bereplaced by alternative elements that provide substantially similarfunctionality or otherwise provide adequate operation. This includes,for example, presently known alternative elements, such as those thatmight be currently known to one skilled in the art, and alternativeelements that may be developed in the future, such as those that oneskilled in the art might, upon development, recognize as an alternative.Further, the disclosed embodiments include a plurality of features thatare described in concert and that might cooperatively provide acollection of benefits. The present invention is not limited to onlythose embodiments that include all of these features or that provide allof the stated benefits, except to the extent otherwise expressly setforth in the issued claims.

The embodiments of the invention in which an exclusive property or privilege is claimed are defined as follows:
 1. A method of communicating data between a remote device and an inductive power supply using keyed modulation, said method comprising: receiving inductive power via a variable frequency power transmission signal from the inductive power supply; encoding a data stream of bits into a fixed frequency communication signal, wherein the data stream is represented in the fixed frequency communication signal by one or more high signal levels and one or more low signal levels; modulating the fixed frequency communication signal on the variable frequency power transmission signal according to first and second respective signal level modulation frequencies by toggling a load of a communication transmitter according to the first respective signal level modulation frequency for each of the one or more high signal levels of the fixed frequency communication signal and by toggling the load of the communication transmitter according to the second respective signal level modulation frequency for each of the one or more low signal levels of the fixed frequency communication signal, wherein each bit of the data stream is represented by a plurality of amplitude modulations generated by toggling the load between a high state and a low state according to at least one of the first and second respective signal level modulation frequencies, wherein the load is toggled on the variable frequency power transmission signal; and varying the first and second respective modulation frequencies based on variations in a frequency of the variable frequency power transmission signal, wherein a number of the plurality of amplitude modulations associated with each bit of the data stream varies based on changes in the first and second respective signal level modulation frequencies.
 2. The method as claimed in claim 1 wherein the communication transmitter is configured to modulate the load, and wherein said modulating includes toggling the load in the communication transmitter at a modulation rate based on one of the first and second respective signal level modulation frequencies, wherein the modulation rate is a fraction of the variable frequency power transmission signal.
 3. The method as claimed in claim 2 wherein the modulation rate is one-half the variable frequency power transmission signal.
 4. The method as claimed in claim 1 wherein said modulating includes producing a modulator control signal formed by exclusive OR-ing (a) a modulation clock that is operating at an even integer fraction of the variable frequency power transmission signal and (b) the fixed frequency communication signal, wherein the variable frequency power transmission signal is modulated according to said modulator control signal.
 5. The method as claimed in claim 1 wherein the communication transmitter is configured to modulate the load at a modulation rate based on one of the first and second respective signal level modulation frequencies, wherein the modulation rate is phase locked at a harmonic of the variable frequency power transmission signal.
 6. The method as claimed in claim 5 wherein the modulation rate is four times the variable frequency power transmission signal.
 7. The method as claimed in claim 1 wherein a high bit of the data stream is communicated by modulating at the first respective signal level modulation frequency, and wherein a low bit of the data stream is communicated by modulating at the second respective signal level modulation frequency.
 8. The method as claimed in claim 7 wherein the first respective signal level modulation frequency is one-eighth of the variable frequency power transmission signal, and the second respective signal level modulation frequency is one-tenth of the variable frequency power transmission signal, wherein the first respective signal level modulation frequency and the second respective signal level modulation frequency vary based on variations in the frequency of the variable frequency power transmission signal, whereby the first respective signal level modulation frequency and the second respective signal level modulation frequency increase as the frequency of the variable frequency power transmission signal increases.
 9. The method as claimed in claim 1 further comprising decoding the data stream from the variable frequency power transmission signal by passing the variable frequency power transmission signal through a frequency discriminator.
 10. The method as claimed in claim 1 wherein the load includes at least one of a resistive element, a capacitive element, and an inductive element.
 11. The method as claimed in claim 1 wherein the communication transmitter modulates the fixed frequency communication signal on the variable frequency power transmission signal such that communication inversion is avoided.
 12. A remote device for receiving power from an inductive power supply and for communicating data from the remote device to the inductive power supply, said remote device comprising: a power receiver adapted to receive a variable frequency power transmission signal; a controller adapted to encode a data stream of bits into a fixed frequency communication signal, wherein the data stream is represented in the fixed frequency communication signal by one or more high signal levels and one or more low signal levels, the controller adapted to generate first and second respective signal level modulation frequencies a based on a frequency of the variable frequency power transmission signal, wherein the first and second respective signal level modulation frequencies vary based on variations in the frequency of the variable power transmission signal; and a communication transmitter adapted to selectively toggle a load between a high state and a low state for modulating the fixed frequency communication signal on the variable frequency power transmission signal, wherein the load is selectively toggled between the high and low states according to the first respective signal level modulation frequency for each of the one or more high signal levels of the fixed frequency communication signal, and wherein the load is selectively toggled between the high and low states according to the second respective signal level modulation frequency for each of the one or more low signal levels of the fixed frequency communication signal; wherein each bit of the data stream is represented by a plurality of amplitude modulations generated by toggling the load between the high and low states on the variable frequency power transmission signal according to at least one of the first and second respective signal level modulation frequencies, wherein a number of the plurality of amplitude modulations associated with each bit of the data stream varies based on changes in the first and second modulation respective signal level modulation frequencies.
 13. The remote device as claimed in claim 12 wherein the communication transmitter is configured to modulate the load, and wherein said modulating includes toggling the load in the communication transmitter at a modulation rate corresponding to one of the first and second respective signal level modulation frequencies, wherein the modulation rate is a fraction of the variable frequency power transmission signal.
 14. The remote device as claimed in claim 13 wherein the modulation rate is one-half the variable frequency power transmission signal.
 15. The remote device as claimed in claim 12 wherein the modulating includes producing a modulator control signal formed by exclusive OR-ing (a) a modulation clock that is operating at one of the first and second respective signal level modulation frequencies and at an even integer fraction of the variable frequency power transmission signal and (b) the fixed frequency communication signal, wherein the variable frequency power transmission signal is modulated according to said modulator control signal.
 16. The remote device as claimed in claim 12 wherein the communication transmitter is configured to modulate the load at a modulation rate that is phase locked at a harmonic of the variable frequency power transmission signal.
 17. The remote device as claimed in claim 16 wherein the modulation rate is four times the variable frequency power transmission signal.
 18. The remote device as claimed in claim 12 wherein a high bit of the data stream is communicated by modulating at the first respective signal level modulation frequency, and wherein a low bit of the data stream is communicated by modulating at the second respective signal level modulation frequency.
 19. The remote device as claimed in claim 18 wherein the first respective signal level modulation frequency is one-eighth of the variable frequency power transmission signal, and the second respective signal level modulation frequency is one-tenth of the variable frequency power transmission signal, wherein the first respective signal level modulation frequency and the second respective signal level modulation frequency vary based on variations in the frequency of the variable frequency power transmission signal, whereby the first respective signal level modulation frequency and the second respective signal level modulation frequency increase as the frequency of the variable frequency power transmission signal increases.
 20. The remote device as claimed in claim 12 wherein the load includes at least one of a resistive element, a capacitive element, and an inductive element.
 21. The remote device as claimed in claim 12 wherein the communication transmitter modulates the fixed frequency communication signal on the variable frequency power transmission signal such that communication inversion is avoided.
 22. A system for transferring power to a remote device from an inductive power supply and for communicating data between the remote device and the inductive power supply using keyed modulation, said system comprising: a power receiver adapted to receive a variable frequency power transmission signal; a controller for encoding a data stream of bits in to a fixed frequency communication signal, wherein the data stream is represented in the fixed frequency communication signal by one or more high signal levels and one or more low signal levels, the controller adapted to generate first and second respective signal level modulation frequencies based on a frequency of the variable frequency power transmission signal, wherein the first and second respective signal level modulation frequencies vary based on variations in the frequency of the variable power transmission signal; and a communication transmitter selectively toggling a load between a high state and a low state for modulating the fixed frequency communication signal on the variable frequency power transmission signal, wherein the load is selectively toggled between the high and low states according to the first respective signal level modulation frequency for each of the one or more high signal levels of the fixed frequency communication signal, and wherein the load is selectively toggled between the high and low states according to the second respective signal level modulation frequency for each of the one or more low signal levels of the fixed frequency communication signal; wherein each bit of the data stream is represented by a plurality of amplitude modulations generated by toggling the load between the high and low states on the variable frequency power transmission signal according to at least one of the first and second respective signal level modulation frequencies, wherein a number of the plurality of amplitude modulations associated with each bit of the data stream varies based on changes in the first and second respective signal level modulation frequencies.
 23. The system as claimed in claim 22 wherein said remote device includes said communication transmitter such that said remote device communicates the data to the inductive power supply using the keyed modulation, and wherein said inductive power supply includes detector circuitry for decoding the data stream from the variable frequency power transmission signal.
 24. The system as claimed in claim 22 wherein said inductive power supply includes said communication transmitter such that said inductive power supply communicates the data to the remote device using said keyed modulation, and wherein said remote device includes detector circuitry for decoding the data stream from the variable frequency power transmission signal.
 25. The system as claimed in claim 22 wherein said communication transmitter is configured to modulate said load, and wherein the modulating includes toggling the load in the communication transmitter at a modulation rate corresponding to one of the first and second respective signal level modulation frequencies, wherein the modulation rate is a fraction of the variable frequency power transmission signal.
 26. The system as claimed in claim 25 wherein the modulation rate is one-half the variable frequency power transmission signal.
 27. The system as claimed in claim 22 wherein the modulating includes producing a modulator control signal formed by exclusive OR-ing (a) a modulation clock that is operating at one of the first and second respective signal level modulation frequencies and at an even integer fraction of the variable frequency power transmission signal and (b) the fixed frequency communication signal, wherein the variable frequency power transmission signal is modulated according to said modulator control signal.
 28. The system as claimed in claim 23 wherein the communication transmitter is configured to modulate the load at a modulation rate corresponding to one of the first and second respective signal level modulation frequencies, wherein the modulation rate is phase locked at a harmonic of the variable frequency power transmission signal.
 29. The system as claimed in claim 28 wherein the modulation rate is four times the variable frequency power transmission signal.
 30. The system as claimed in claim 22 wherein a high bit of the data stream is communicated by modulating at the first respective signal level modulation frequency, and wherein a low bit of the data stream is communicated by modulating at the second respective signal level modulation frequency.
 31. The system as claimed in claim 30 wherein the first respective signal level modulation frequency is one-eighth of the variable frequency power transmission signal, and the second respective signal level modulation frequency is one-tenth of the variable frequency power transmission signal, wherein the first respective signal level modulation frequency and the second respective signal level modulation frequency vary based on variations in the frequency of the variable frequency power transmission signal, whereby the first respective signal level modulation frequency and the second respective signal level modulation frequency increase as the frequency of the variable frequency power transmission signal increases.
 32. The system as claimed in claim 22 further comprising an inductive power supply controller configured to decode the data stream from the variable frequency power transmission signal by passing the variable frequency power transmission signal through a frequency discriminator.
 33. The system as claimed in claim 22 wherein the load includes at least one of a resistive element, a capacitive element, and an inductive element.
 34. The system as claimed in claim 22 wherein the communication transmitter modulates the fixed frequency communication signal on the variable frequency power transmission signal such that communication inversion is avoided.
 35. An inductive power supply for transferring inductive power to a remote device and for receiving data via backscatter modulation from the remote device, said inductive power supply comprising: a power transmitter for transmitting a variable frequency power transmission signal, the power transmitter including a primary configured to generate the variable frequency power transmission signal based on a power supply signal; switching circuitry operably coupled to the power transmitter, the switching circuity for supplying the power supply signal to the primary at a power transmission frequency such that the variable frequency power transmission signal is transmitted from the primary substantially at the power transmission frequency; a controller operably coupled to the switching circuitry, the controller configured to control the power transmission frequency of the switching circuitry, the controller configured to vary the power transmission frequency to vary the variable frequency power transmission signal; and a wireless communication receiver configured to detect modulation on the variable frequency power transmission signal, wherein the inductive power supply decodes a data stream from the variable frequency power transmission signal, wherein each bit of the data stream is represented by a plurality of amplitude modulations generated by toggling a load between a high state and a low state on the variable frequency power transmission signal, and wherein a number of the plurality of amplitude modulations associated with each bit of the data stream varies based on variations in the variable frequency transmission signal; and wherein the data stream is encoded into a fixed frequency communication signal that is modulated on the variable frequency power transmission signal according to first and second respective signal modulation frequencies, wherein the load is selectively toggled between the high and low states according to the first respective signal level modulation frequency for each of one or more high signal levels of the fixed frequency communication signal, and wherein the load is selectively toggled between the high and low states according to the second respective signal level modulation frequency for each of one or more low signal levels of the fixed frequency communication signal.
 36. The inductive power supply as claimed in claim 35 wherein the fixed frequency communication signal is a bi-phase encoded representation of the data stream.
 37. The inductive power supply as claimed in claim 35 wherein the inductive power supply decodes the data stream from the variable frequency power transmission signal by passing the variable frequency power transmission signal through a frequency discriminator. 